System and method of modelling capacitance of on-chip coplanar transmission line structures over a substrate

ABSTRACT

Methods, systems and apparatus for modeling capacitance for a structure comprising a pair of long conductors surrounded by a dielectric material and supported by a substrate. In particular, the structure may be on-chip coplanar transmission lines over a conductive substrate operated at very high frequencies, such that the substrate behaves as a perfect dielectric. It is assumed that the surrounding dielectric material is a first dielectric with a first permittivity (e 1 ) and the substrate is a second dielectric with a second permittivity (e 2 ). A method models the capacitance (C 1 ) for values of the first and second permittivity (e 1 , e 2 ) based on known capacitance (C 2 ) computed for a basis structure with the same first permittivity (e 1 ) and a different second permittivity (e 2 ). Extrapolation or interpolation formulae are suggested to model the sought capacitance (C 1 ) through one or more known capacitances (C 2 ).

TECHNICAL FIELD

The present invention relates to the field of modeling capacitance. In particular, the invention relates to modeling capacitance in integrated circuits; specifically, modeling capacitance of on-chip coplanar transmission lines over a substrate.

BACKGROUND OF THE INVENTION

Models of on-chip coplanar transmission lines over a substrate have been developed as a part of the “T-lines set” which is the core of an interconnect-aware design and modeling methodology, enabling high predictability of the critical interconnect behavior. Further information is provided in references: Goren, D. et al., “An Interconnect-Aware Methodology for Analog and Mixed Signal Design, Based on High Bandwidth (Over 40 GHz) On-chip Transmission Line Approach” IEEE DATE '02 Conference, Paris March 2002, pp. 804-811 and Goren, D. et al., “On-chip Interconnect-Aware Design and Modeling Methodology, Based on High Bandwidth Transmission Line Devices”, IEEE DAC '03 Conference, CA, June 2003, pp. 724-727.

Currently supported production level coplanar transmission lines include single wire and two coupled wires as shown in the cross-sections of FIGS. 1A and 1B. The structures are symmetrical and include grounded side shield lines. The wires are close to each other and “thick” i.e. their width, the metal layer thickness and the gaps between the wires are of the same order. The metal wires are surrounded in cross-section by a surrounding material which is a dielectric, for example, an oxide dielectric. The surrounding material is supported by a substrate, for example, a silicon substrate.

As follows from hardware measurements and numerical studies of wideband behavior of on-chip coplanar transmission lines over a substrate which is conductive, a line capacitance per unit length strongly decreases with the frequency growth of the current in the wires. Although a lower substrate resistivity causes a slower capacitance decrease, the C(f) curves for different substrate resistance values tend to the same asymptotic values: a low frequency limit C₀ and a high frequency limit C_(¥).

The surrounding material in which the wires are provided is assumed to be a dielectric which is a perfect insulator. The substrate is a material which has a finite electrical conductivity in addition to dielectric properties.

The permittivity e₁ of the dielectric surrounding material does not depend on frequency, while the permittivity e₂ of a conductive substrate is a complex quantity depending on the frequency f, such that at f=0 and at f=¥ it assumes real values. The substrate acts as a perfect conductor at low frequencies and as an ideal dielectric at very high frequencies. As a result of the frequency dependence of the substrate, the capacitance of the conductors is a function of frequency. A solution for arbitrary real permittivities e₁ and e₂ includes both C₀ and C_(¥) as specific cases. Accurate computation of these asymptotical values has critical impact on the accuracy of a transmission line model in the whole bandwidth of interest.

There are several techniques for computing C₀ which account for “thick” wires and result in explicit expressions yielding reasonable accuracy. However, the previous approaches to calculating C_(¥) either have used purely numeric approach, or have been based on very rough assumptions (such as assuming zero thickness of the wires) leading to large and unpredictable errors.

An Electro-Magnetic (EM) solver provides a numeric solution of Maxwell's equations for certain specific cases, characterized by geometry (2D or 3D) and frequency range (quasi-static or full wave solution method is chosen based on the required frequency range per geometry). The output includes electric and/or magnetic field distributions, and integral parameters (capacitance, inductance, S-parameters, impedance), etc. In any case, for each set of input parameters, the solution process includes a definition of the solution domain, dividing it into a mesh of N elementary cells, building and solving a system of k*N algebraic linear equations, k³1. N has a variable order of magnitude, 10²-10⁶, depending on geometry and required accuracy.

The advantage of EM solvers is their high accuracy. Their disadvantages are very large computation time and memory usage, and extremely difficult integration with other tools, which prevents their usage in various solution flows/environments, e.g. chip design and simulation flow. Therefore, EM solvers are used mostly as stand-alone tools for verifying or fitting of other solutions which are integrated within the desired design flow or environment.

The aim of the present invention is to yield accurate explicit expressions for high frequency capacitance C_(¥) which account for “thick” rectangular wires. In particular, a method of modeling capacitance is provided for a structure such as on-chip coplanar transmission lines over a conductive substrate operated at very high frequencies such that the substrate behaves as a perfect dielectric.

There are specific cases of the relationship between the permittivities e₁ and e₂ of the surrounding material and the substrate in which capacitance is much easier to calculate. The core idea is to calculate the capacitance of a pair of long conductors for certain values of permittivity e₁ and e₂, based on a known capacitance computed for the same value of the permittivity e₁ of the surrounding material and a different value of the permittivity e₂ of the substrate.

The present invention uses the fact that at very high frequencies, the substrate behaves as a dielectric. Therefore, formulae can be suggested for computing the capacitance in the case of two different dielectrics as the surrounding material and the substrate. These formulae can then be used for modeling the desired high frequency capacitance.

The description presents a semi-analytical technique for modeling capacitance of on-chip coplanar transmission lines over a substrate. The focus is put on developing expressions for high frequency capacitance which yield reasonable accuracy. The technique is based on the two-dimensional approach and results in accurate and efficient expressions accounting for frequency dependent behavior of the substrate, as well as for actual transmission lines geometry.

SUMMARY OF THE INVENTION

According to a first aspect of the present invention there is provided a method of modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by a substrate, wherein the capacitance of the conductors is a function of current frequency, the method comprising: assuming the surrounding dielectric material is a first dielectric with a first permittivity (e₁) and the substrate is a second dielectric with a second permittivity (e₂); modeling the capacitance (C₁) for values of the first and second permittivity (e₁, e₂) based on known capacitance (C₂) computed for a basis structure with the same first permittivity (e₁) and a different second permittivity (e₂). The known capacitance is advantageously computed for a basis structure with the same cross-section geometry as the subject structure.

According to a second aspect of the present invention there is provided a method of modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by a substrate, wherein the capacitance of the conductors is a function of current frequency, wherein the method includes the steps of: assuming the surrounding dielectric material is a first dielectric with a first permittivity (e₁) and the substrate is a second dielectric with a second permittivity (e₂); determining the reflection coefficient for the subject structure and the basis structure based on the first and second permittivities (e₁, e₂); determining the mirror capacitance for the basis structure; and computing the capacitance of the subject structure based on the reflection coefficients and the mirror capacitance.

According to a third aspect of the present invention there is provided an integrated circuit design system comprising means for modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by substrate.

According to another aspect of the present invention there is provided a computer program product stored on a computer readable storage medium, comprising computer readable program code means for modeling capacitance for a subject structure.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will now be described, by way of examples only, with reference to the accompanying drawings in which:

FIGS. 1A and 1B are cross-sections of coplanar transmission lines as known in the prior art; and

FIGS. 2A and 2B are cross-sections of a structure defining a coordinate system in accordance with the present application;

FIGS. 3A to 3C are flow diagrams showing the steps of modeling capacitance in accordance with the present invention;

FIG. 4 is a schematic representation of an integrated circuit design system for implementing a design process including modeling capacitance in accordance with the present invention.

DESCRIPTION OF THE INVENTION

The present invention provides methods, apparatus and systems for modeling capacitance in integrated circuits; specifically, modeling capacitance of on-chip coplanar transmission lines over a substrate. According to a first aspect of the present invention there is provided a method of modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by a substrate, wherein the capacitance of the conductors is a function of current frequency, the method comprising: assuming the surrounding dielectric material is a first dielectric with a first permittivity (e₁) and the substrate is a second dielectric with a second permittivity (e₂); modeling the capacitance (C₁) for values of the first and second permittivity (e₁, e₂) based on known capacitance (C₂) computed for a basis structure with the same first permittivity (e₁) and a different second permittivity (e₂). The known capacitance is advantageously computed for a basis structure with the same cross-section geometry as the subject structure.

The surrounding dielectric material is advantageously an insulating material and the substrate advantageously behaves as a conductor at low frequencies and as a dielectric at very high frequencies. Therefore, in an assumed model, the first permittivity does not depend on frequency and the second permittivity does depend on frequency. This method of modeling capacitance is particularly applicable to modeling high frequency capacitance as a substrate may act as a dielectric at high frequencies. The capacitance of the subject structure may be modeled from the capacitance of one or more basis structures using an extrapolation or interpolation formula.

The input information for the method may include: the first permittivity e₁ of the first dielectric of both the subject structure and the basis structure, the second permittivity e₂ of the second dielectric of the basis structure, and the cross-sectional geometry of the subject or basis structure. The known capacitance may be provided by one of an Electro-Magnetic solver, an analytical formula, a quasi-analytical formula, and lookup tables.

In an example embodiment, the dielectric is an oxide dielectric and the substrate is a silicon substrate. The pair of long conductors may each have a width and metal layer thickness of the same order. The pair of long conductors may be spaced apart by a distance of the same order as the width and thickness of the conductors. The cross-section of each of the long conductors may be constant along the length. Each of the long conductors may be formed of a single conductor or more than one parallel conductor. Advantageously, the structure comprises on-chip coplanar transmission lines over a conductive substrate.

According to a second aspect of the present invention there is provided a method of modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by a substrate, wherein the capacitance of the conductors is a function of current frequency, wherein the method includes the steps of: assuming the surrounding dielectric material is a first dielectric with a first permittivity (e₁) and the substrate is a second dielectric with a second permittivity (e₂); determining the reflection coefficient for the subject structure and the basis structure based on the first and second permittivities (e₁, e₂); determining the mirror capacitance for the basis structure; and computing the capacitance of the subject structure based on the reflection coefficients and the mirror capacitance.

The mirror capacitance may be computed using the charge density function of the basis structure. Alternatively, the mirror capacitance may be the averaged mirror capacitance based on a uniform charge density approximation for the basis structure.

The method may include the following steps: defining: C₁ as the capacitance for the subject structure; C₂ as the capacitance for the basis structure; e₂₁ as the substrate permittivity e₂ of the subject structure; and e₂₂ as the substrate permittivity e₂ of the basis structure; determining the reflection coefficient, k_(i)=(e₁−e_(2i))/(e₁+e_(2i)) for i=1,2; determining if the charge density function s(X,k₂) for the basis solution is available, if it is available, computing the “mirror capacitance” C*(k₂)=C*(X₁,X₂,k₂) which is fully defined by s(X,k₂), and if the charge density function for the basis solution is not available, computing the “averaged mirror capacitance” C*(k₂)=C*_(avg)(k₂) based on the uniform charge density approximation; and computing the capacitance C₁ using the extrapolation formula: C₁=1/(1/C₂+(k₁−k₂)/C*(k₂)).

According to a third aspect of the present invention there is provided an integrated circuit design system comprising means for modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by substrate, assuming the surrounding dielectric material is a first dielectric with a first permittivity (e₁) and the substrate is a second dielectric with a second permittivity (e₂), the system comprising: means for modeling the capacitance (C₁) for values of the first and second permittivity (e₁, e₂) based on known capacitance (C₂) computed for a basis structure with the same first permittivity (e₁) and a different second permittivity (e₂).

According to a fourth aspect of the present invention there is provided an integrated circuit design system comprising means for modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by substrate, assuming the surrounding dielectric material is a first dielectric with a first permittivity (e₁) and the substrate is a second dielectric with a second permittivity (e₂), the system comprising: means for determining the reflection coefficient for the subject structure and a basis structure based on the first and second permittivities (e₁, e₂) of the subject structure and the basis structure; means for determining the mirror capacitance for the basis structure; and means for computing the capacitance of the subject structure based on the reflection coefficients and the mirror capacitance.

The integrated circuit design system may comprise: means for determining the reflection coefficient, k_(i)=(e₁−e_(2i))/(e₁+e_(2i)) for i=1,2; means for determining if the charge density function s(X,k₂) for the basis solution is available, if it is available, means for computing the “mirror capacitance” C*(k₂)=C*(X₁,X₂,k₂) which is fully defined by s(X,k₂), and if the charge density function for the basis solution is not available, means for computing the “averaged mirror capacitance” C*(k₂)=C*_(avg)(k₂) based on the uniform charge density approximation; and means for computing the capacitance C₁ using the extrapolation formula: C₁=1/(1/C₂+(k_(i)−k₂)/C*(k₂)); wherein C₁ is the capacitance for the subject structure; C₂ is the capacitance for the basis structure; e₂₁ is the substrate permittivity e₂ of the subject structure; and e₂₂ is the substrate permittivity e₂ of the basis structure.

According to the fifth aspect of the present invention there is provided a computer program product stored on a computer readable storage medium, comprising computer readable program code means for modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by a substrate, assuming the surrounding dielectric material is a first dielectric with a first permittivity (e₁) and the substrate is a second dielectric with a second permittivity (e₂), comprising code means for performing the modeling the capacitance (C₁) for values of the first and second permittivity (e₁, e₂) based on known capacitance (C₂) computed for a basis structure with the same first permittivity (e₁) and a different second permittivity (e₂).

According to a sixth aspect of the present invention there is provided a computer program product stored on a computer readable storage medium, comprising computer readable program code means for modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by a substrate, assuming the surrounding dielectric material is a first dielectric with a first permittivity (e₁) and the substrate is a second dielectric with a second permittivity (e₂), comprising code means for performing the steps of: determining the reflection coefficient for the subject structure and the basis structure based on the first and second permittivities (e₁, e₂); determining the mirror capacitance for the basis structure; and computing the capacitance of the subject structure based on the reflection coefficients and the mirror capacitance.

The computer program product may comprise computer readable program code means for performing the steps of: determining the reflection coefficient, k_(i)=(e₁−e_(2i))/(e₁+e_(2i)) for i=1,2; determining if the charge density function s(X,k₂) for the basis solution is available, if it is available, computing the “mirror capacitance” C*(k₂)=C*(X₁,X₂,k₂) which is fully defined by s(X,k₂), and if the charge density function for the basis solution is not available, computing the “averaged mirror capacitance” C*(k₂)=C*_(avg)(k₂) based on the uniform charge density approximation; and computing the capacitance C₁ using the extrapolation formula: C₁=1/(1/C₂+(k_(i)−k₂)/C*(k₂)); wherein C₁ is the capacitance for the subject structure; C₂ is the capacitance for the basis structure; e₂₁ is the substrate permittivity e₂ of the subject structure; and e₂₂ is the substrate permittivity e₂ of the basis structure.

FIGS. 1A and 1B show the cross-sections of currently supported production level coplanar transmission lines. FIG. 1A shows a first example 100 with a single wire 101. FIG. 1B shows a second example with two coupled wires 111, 112. The structures are symmetrical and include grounded side shield lines 103, 104, 113, 114.

The signal wires 101, 111, 112, and the grounded side shield lines 103, 104, 113, 114 are supported in a surrounding material 105, 115 which is a dielectric, for example, an oxide dielectric, and are based on a substrate 106, 116, for example, a silicon substrate.

The wires are “thick” i.e. their width and the metal layer thickness are of the same order. The transmission line structures are “long” in a sense that they are featured by a small ratio of cross-section characteristic dimension versus wire length, usually less than 2%.

For such structures, calculation of all electric characteristics can be performed per unit length using a two-dimensional approach. A line capacitance per unit length of on-chip coplanar transmission lines over a conductive substrate strongly decreases with the frequency growth. This is known from hardware measurements and numerical studies of wideband behavior. Although a lower substrate resistivity causes a slower capacitance decrease, the C(f) curves for different substrate resistance values tend to the same asymptotic values: the low frequency limit C₀ and the high frequency limit C₈. Accurate computation of these asymptotical values has critical impact on the accuracy of a transmission line model in the whole bandwidth of interest. There are several techniques for computing the low frequency limit of capacitance C₀ which account for “thick” wires and result in explicit expressions yielding reasonable accuracy. This description focuses on developing accurate explicit expressions for the high frequency limit of capacitance C₈ which account for “thick” wires.

A pair of long conductors is considered located in a uniform lossless dielectric over a conductive substrate. The conductors do not touch the planar boundary 108, 118 between the dielectric and the substrate. It is assumed that the substrate is “grounded at infinity”, namely that far away from the conductors the potential of the substrate is zero. This condition is what is provided in practical designs, assuming the mandatory existence of substrate ground contacts, though in many practical cases there is no back ground metallization contact to the substrate.

The conductors' cross-sections can be of any shape, provided that it is constant along the wires, in particular, each conductor may be composed of several parallel wires.

The transmission line structures shown in FIGS. 1A and 1B are considered as examples. In case of a single coplanar transmission line 100 as shown in FIG. 1A, a pair of long conductors is provided in which the first conductor is the signal wire S 101 and the second conductor is the union of the side shield wires G 103, 104. Referring to FIG. 1B, for coupled coplanar transmission line operated in common mode, a pair of long conductors is provided in which the first conductor is the union of signal wires S₁ 111 and S₂ 112 and the second conductor is the union of two side shield wires G 113, 114. While in single mode, the first conductor is one of the signal wires 111, 112, and the second conductor is the union of another signal wire 112, 111 and both side shield wires 113, 114.

Both the surrounding dielectric material and the substrate should be spatially uniform i.e. both the permittivity e₁ of the surrounding dielectric material and the permittivity e₂ of the substrate do not depend on coordinates. The permittivity e₁ of the surrounding material does not depend on frequency. However, the permittivity e₂ of the substrate does depend on frequency f, such that at low frequencies the substrate behaves as a perfect conductor: e₂(f=0)® 8, and at very high frequencies—as an ideal dielectric: 8>e₂(f® 8)³e₁. This frequency dependence of e₂ usually follows from the substrate having a finite electrical conductivity in addition to its dielectric properties. However, the surrounding dielectric material is assumed to be a perfect insulator.

In an example embodiment, the surrounding dielectric material may be an oxide dielectric such as silicon oxide which is a good example for such a practically zero conductivity material to be used as the dielectric. The conductive substrate may be, for example, a silicon substrate.

As stated above, the permittivity e₁ of the surrounding dielectric material does not depend on frequency, while the permittivity e₂(f) of the substrate is a complex quantity depending on the frequency f, such that at f=0 and at f® 8 it assumes real values: ε₁<ε₂=ε₂(f→∞)<ε₂(f=0)→∞

In case of silicon oxide dielectric over silicon substrate, e₁˜4, e₂(f® ¥)˜12.

Thus, at low frequencies the substrate behaves as a perfect conductor, and at very high frequencies—as an ideal dielectric. As a result, the capacitance of the conductors is a function of frequency. Both asymptotic values of the capacitance, C₀ and C₈, can be calculated assuming that the electric charge is distributed on the surface of the wires, i.e. using an electrostatic approach.

The transition from “conducting behavior” to “dielectric behavior” can be estimated by a transition frequency usually referred to as “relaxation frequency”. freql=1/(2pe ₀ e _(sub) r _(sub)) where: r_(sub) is the resistivity of the substrate (typically 1 to 15 Ohm cm);

-   -   e_(sub) is the relative permittivity of the substrate (for         silicon, e_(sub)=11.9);     -   e₀ is the permittivity of a vacuum.

For example, for CMOS, r_(sub)=1-2 Ohm cm which gives a relaxation frequency,

-   freql in the order of 100 GHz, for SOI and SiGe, r_(sub)=10-15 Ohm     cm which gives a relaxation frequency, freql in the order of 10 GHz.

The goal of the described method is to evaluate the capacitance for arbitrary real e₁=e₂ which includes both C₀ and C₈ as specific cases. A two-dimensional approach is used, i.e. all the electric field characteristics refer to an arbitrary cross-section of the structure which is perpendicular to the conductors' length.

In this approach, as the substrate behaves as a dielectric at very high frequencies, the surrounding material and the substrate are modeled as two different dielectrics, a first dielectric with permittivity e₁ and a second dielectric with permittivity e₂.

In an arbitrary cross-section plane, a coordinate system YZ is introduced as shown in FIG. 2A. A cross-section 200 of a structure is shown. The structure is formed of a first dielectric e₁ 205 and second dielectric e₂ 206 with a pair of long conductors in the form of wires P1 203 and P2 204 disposed in the first dielectric e₁ 205. An axis Y 211 is defined lying within the planar e₁/e₂ boundary 202 between the first dielectric e₁ 205 and the second dielectric e₂ 206. An axis Z 212 is defined perpendicular to the boundary 202. The origin of the coordinate system is chosen to simplify the calculations.

The calculations of the described method are presented in three stages.

-   1. Derive the electrostatic potential j induced by an infinitely     thin and long charged wire located in the half-space filled with     dielectric e₁ (further referred to as “upper half-space”) near the     planar boundary with the half-space filled with dielectric e₂ ³e₁     (“lower half-space”). -   2. Derive the electrostatic potentials V₁ and V₂ of two arbitrarily     shaped infinitely long conductors bearing charge Q and −Q per unit     length. -   3. The capacitance per unit length is C=Q/|V₁−V₂|.     Stage 1—Electrostatic Potential Induced by a Thin Charged Wire     Parallel to a Planar Boundary Between Two Dielectrics

Denote by s the wire charge per unit length, and by j₁, j₂ potential in the dielectrics e₁ and e₂, respectively. Specify the origin of the coordinate system YZ on the e₁/e₂ boundary exactly below the wire, so that the axis Z crosses the wire at the point X₀(y=0, z=z₀) as shown in FIG. 2B.

Within each of the dielectrics, e₁ and e₂ satisfy the Poisson equation ∇²φ_(i)=σ_(i)/ε_(i) , i=1,2.  (1) s₁, s₂ are charge densities in the dielectrics e₁, e₂, defined as s₁=s*d(X−X₀) where d(X−X₀) is the 2-D Dirac delta-function, and s₂=0. Therefore, in the infinitely close vicinity of the point X₀, the potential is 0 j ₁ ⁰ =s/(2

e ₁)ln(R ₀ /R) where R is the distance between the point X₀ and an arbitrary point X close to X₀, R₀ is an arbitrary constant distance. The potentials j₁, j₂ can be expressed as follows: j ₁ =j ₁ ⁰ +j ₁ *, Ñ ² j ₁*=0, Ñ²j₂=0,  (2) the functions j₁* and j₂ having no singularities in respective domains.

The potential in any cross-section point X is a function of only two independent distances: the distance R between X and X₀, and the distance z between X and the e₁/e₂ boundary. For the sake of symmetry, instead of z, R* is used—the distance between X and X₀* (y=0, z=−z₀), which is symmetrical to X₀ relative to the e₁/e₂ boundary (see FIG. 2B).

On the planar e₁/e₂ boundary (z=0, or R=R*), the potential should meet the following boundary conditions: j₁=j₂, e ₁ j ₁ /z=e ₂ j ₂ /z.  (3)

The potential in the dielectrics e₁ and e₂ which satisfies both the Equation 1 and the boundary conditions Equation 3, is j=j₁ for z>0, j=j₂ for z<0, j ₁ =s/(2

e ₁)(ln(R ₀ /R)+kln(R ₀ /R*)), j ₂ =s/(2

e ₂)(1−k)ln(R ₀ /R),  (4) where k=(e ₁ −e ₂)/(e ₁ +e ₂), R=Ö(y ²+(z−z ₀)²), R*=Ö(y ²+(z+z ₀)²).

In order to compute capacitance of the wires located in the upper half-space, only the potential j₁ is needed. As can be seen from Equation 4, j₁ will not change if the dielectric e₂ in the low half-space is replaced by e₁, and a second wire bearing charge ks per unit length is inserted, which is the mirror image of the initial wire in the planar e₁/e₂ boundary. (It should be noted that the described operation changes the potential in the lower half-space.)

The important specific cases of the solution given by Equation 4 are:

-   -   k=0         e₁=e₂—the wire is located in the space filled with uniform         dielectric e₁.     -   k=−1         e₂® 8—the wire is located in the upper half-space filled with         dielectric e₁, while the lower half-space is filled with perfect         conductor. This case corresponds to the behavior of the         substrate at the low frequency limit.

For the high frequency limit, e₁<e₂<8

−1<k<0.

Stage 2—Electrostatic Potential Induced by Two Infinitely Lone Oppositely Charged Conductors Parallel to a Planar Boundary Between Different Dielectrics.

Consider a pair of long conductors located in the upper half-space filled with dielectric e₁. Let e₂₁ and e₂₂ be different values of e₂, k₁=k(e₂₁), k₂=k(e₂₂), hence, 0³ k₁,k₂ ³−1. For any physical characteristic F(x, k) defined on surface of the conductors, the following notations will be used (i=1,2): dF(x, k ₁ , k ₂)=F(x, k ₁)−F(x, k ₂), F(x,k)=F _(i)(x,k) for xÎP _(i), ·F(x ₁ , x ₂ , k)=F ₁(x ₁ , k)−F ₂(x ₂ , k), x _(i) ÎP _(i), ·F(k)=F ₁(k)−F ₂(k), if F _(i)(x,k)=F _(i)(k) for xÎP _(i).

Denote the conductors' cross-section contours by P₁ and P₂, the perimeter lengths by p₁ and p₂, and the conductors' surface charge density by s(x), so that s(x)=s_(i)(x) for xÎP_(i). The charges of the conductors per unit length are

$\begin{matrix} {{Q_{i} = {\int_{P_{i}}{{\sigma(x)}\ {\mathbb{d}{P(x)}}}}},{i = 1},2.} & (5) \end{matrix}$

The potential induced by the pair of conductors at a point X location

$\begin{matrix} \begin{matrix} {\left. {{V(X)} = {{1/\left( {2{\pi ɛ}_{1}} \right)}{\int_{P_{1}\bigcup P_{2}}{{\sigma(x)}{\varphi_{1}\left( {X,x} \right)}}}}} \right)\ {\mathbb{d}{P(x)}}} \\ {= {{{- 1}/\left( {2{\pi ɛ}_{1}} \right)}{\int_{P_{1}\bigcup P_{2}}{{\sigma(x)}\left( {{\ln\left( {{R\left( {X,x} \right)}/R_{0}} \right)} +} \right.}}}} \\ {\left. {k\mspace{14mu}\ln\left( {{R\left( {X,x^{*}} \right)}/R_{0}} \right)} \right)\ {\mathbb{d}{P(x)}}} \end{matrix} & (6) \end{matrix}$ where the point xÎP₁ÉP₂, x* is the mirror image of the point x in the e₁/e₂ boundary. Stage 3—Capacitance Per Unit Length of Two Arbitrarily Shaped Oppositely Charged Conductors Parallel to a Planar Boundary Between Different Dielectrics.

Consider e₁ as a constant, and e₂ decreasing from 8 to e₁ i.e. k=k(e₂) increasing from −1 to 0, which covers our cases of interest. Then, potential in the upper half-space induced by the pair of conductors with the surface charge density s(x, k) is a function of parameter k.

A relationship between C(k₁) and C(k₂) is sought where k₁=k(e₂₁), k₂=k(e₂₂), e₂₁ and e₂₂ are different values of e₂. An increase of dielectric permittivity in any part of the space causes increase of the capacitance, hence, k ₁ ³ k ₂

e ₂₁ =e ₂₂

C(k ₁)=C(k ₂).

The difference between the values of potential at the point X, dV(X, k₁, k₂), depends on parameters k₁, k₂ explicitly, and also implicitly, as a function of difference between respective charge densities, ds(x, k₁, k₂):

$\begin{matrix} \begin{matrix} {{\delta\;{V\left( {X,k_{1},k_{2}} \right)}} = {{V\left( {X,k_{1}} \right)} - {V\left( {X,k_{2}} \right)}}} \\ {{= {{\left( {k_{1} - k_{2}} \right){V^{*}\left( {X,k_{2}} \right)}} + {V^{\delta}\left( {X,k_{1},{{\delta\sigma}\left( {X,k_{1},k_{2}} \right)}} \right)}}},} \end{matrix} & (7) \\ {{{V^{*}\left( {X,k_{2}} \right)} = {{{- 1}/\left( {2{\pi ɛ}_{1}} \right)}{\int_{P_{1}\bigcup P_{2}}{{\sigma\left( {x,k_{2}} \right)}{\ln\left( {{R\left( {X,x^{*}} \right)}/R_{0}} \right)}\ {\mathbb{d}{P(x)}}}}}},} & (8) \\ {{V^{\delta}\left( {X,k_{1},{{\delta\sigma}\left( {X,k_{1},k_{2}} \right)}} \right)} = {{{- 1}/\left( {2{\pi ɛ}_{1}} \right)}{\int_{P_{1}\bigcup P_{2}}{{{\delta\sigma}\left( {x,k_{1},k_{2}} \right)}\left( {{\ln\left( {{R\left( {X,x} \right)}/R_{0}} \right)} + {k_{1}{\ln\left( {{R\left( {X,x^{*}} \right)}/R_{0}} \right)}}} \right){{\mathbb{d}{P(x)}}.}}}}} & (9) \end{matrix}$

Consider potential difference between the two conductors for k=k₁, ·V(k₁). Since the potential on the surface of each conductor is constant, ·V(k₁) can be measured between any pair of points X₁ÎP₁ and X₂ÎP₂, ·V(k ₁)=·V(X ₁ ,X ₂ ,k ₁).

It follows from Equation 5 that ·V(k₁)>0. Using the Equations 7-9, ·V(k₁) can be represented in the form ·V(k ₁)= ·V(k ₂)+(k ₁ −k ₂)·V*(X ₁ ,X ₂ ,k ₂)+·V ^(d)(X ₁ ,X ₂ ,k ₁ ,ds).  (10)

Note that the charge Q of conductors per unit length does not depend on any pair of values k₁, k₂,

$\begin{matrix} {{{\int_{P_{i}}{{{\delta\sigma}\left( {p,k_{1},k_{2}} \right)}\ {\mathbb{d}{P(x)}}}} = 0},{i = 1},2} & (11) \end{matrix}$

Dividing Equation 10 by Q, gets 1/C(k ₁)=1/C(k ₂)+(k ₁ −k ₂)/C*(X ₁ ,X ₂ , k ₂)+1/C ^(d),  (12) where 1/C(k _(i))=·V(k _(i))/Q, 1/C*(X ₁ ,X ₂ , k ₂)=·V*(X₁ ,X ₂ ,k ₂)/Q, 1/C ^(d =)1/C ^(d)(X ₁ ,X ₂ ,k ₁ , ds)=·V ^(d)(X ₁ ,X ₂ ,k ₁ , ds)/Q.

The term C*(X₁,X₂,k₂) which is referred to as “mirror capacitance” depends on the charge density function s(x,k₂). Once this function is known, both C(k₂) and C*(X₁,X₂,k₂) can be calculated.

An analysis of the term 1/C^(d) yields the following estimation: |1/C ^(d)(X ₁ ,X ₂ ,k ₁ , ds)|<1/(2

e ₁)a(G+|k ₁|2A).

The term A depends on the conductors' cross-sections shape and on their location relative to the e₁/e₂ boundary, A=ln((z ² _(max) +y ² _(max))^(1/2) /z _(min)), z_(min) and z_(max) are the minimal and maximal vertical distance between the conductors' cross-sections contours and the e₁/e₂ boundary, 2*y_(max) is the maximal horizontal distance between the points on the conductors cross-sections contours. The wires do not touch the planar e₁/e₂ boundary, so z_(min)>0. The term G depends on the conductors' cross-sections shape and location of the points X₁, X₂, but does not depend on their location relative to the e₁/e₂ boundary (one can try to find location of X₁, X₂ so that to minimize G). The term a is a maximum relative difference between the conductors' surface charge densities corresponding to different permittivity values of the lower half-space: a=max(a ₁ ,a ₂), a _(i)=(ds _(i))_(max)/(Q/p _(i)), i=1,2.

Dependence of the conductors' capacitance on their surface charge density deviations is weak, if they satisfy Equation 5. The smaller is the difference between k₂ and k₁, the smaller is a and, consequently, 1/C^(d). The computations show that even for k₁=−0.5,

-   k₂=−1, this term is relatively small.

The usefulness of Equation 12 lies in the fact that it enables the calculation of capacitance C(k₁) of a pair of long conductors for certain values of permittivity e₁ and e₂, based on a known capacitance C(k₂) computed for the same value of e₁ and a different value of e₂.

Consider two important cases when C(k₂) is much easier to calculate. Using Equation 12 with k₁=k, k₂=0, C(k) is presented as a deviation from C(−1)—capacitance in case of the whole space filled with dielectric e₁, 1/C(k)Ë1/C(0)+k/C*(X₁,X₂,0), 1/C(k)<1/C(0).  (13)

Similarly, using k₁=k, k₂=−1, C(k) is presented as a deviation from C(−1) which is equal to C₀—capacitance in case of the lower half-space filled with perfect conductor, 1/C(k)Ë1/C(−1)+(k+1)/C*(X₁,X₂,−1), 1/C(k)>1/C(−1).  (14)

If −0.5<k<0, then Equation 13 is more accurate; if −1<k<−0.5, then Equation 14 yields better accuracy. The worst case is k=−0.5. Note that for silicon oxide dielectric over silicon substrate k>>−0.5.

Equation 12 is useful when the charge density function s(x,k₂) is known. However, there are methods that yield C(k₂) in the form from which deriving s(x,k₂) is impossible. An equation similar to Equation 12 is developed which does not require s(x,k₂) to be known.

On the surface of each conductor, potential should be constant, so it can be expressed as its average taken along the cross-section contour:

V_(i)(k) = V_(avg_(i))(k) = −1/(2πɛ₁p_(i))∫_(P_(i))V(X, k)𝕕P(X), i = 1, 2.

The above equation is exact if V(X,k) is the physical potential function. If V(X,k) is an approximate solution, this equation can be used as a condition which this solution should satisfy.

Similarly, define V*_(avg)(k₂) and V^(d) _(avg)(k₁,ds). Eventually, this presentation leads to an equation similar to Equation 12, 1/C(k ₁)=1/C(k₂)+(k ₁ −k ₂)/C* _(avg)(k ₂)+1/C ^(d) _(avg),  (15) where C(k₁) and C(k₂) have the same meaning as in Equation 12, 1/C* _(avg)(k ₂)=·V* _(avg)(k ₂)/Q, 1/C ^(d) _(avg)=1/C ^(d) _(avg)(k ₁ , ds)=·V ^(d) _(avg)(k ₁ , ds)/Q.

The term 1/C^(d) _(avg) (k₁, ds) is small relative to the inverse “averaged mirror capacitance” 1/C*_(avg)(k2), which depends on the charge density s(x,k₂), but the dependence is fairly weak, provided that Equation 5 is satisfied. So, 1/C*_(avg)(k₂) is evaluated using uniform charge density approximation, s ₁(x)=Q/p ₁ , s ₂(x)=−Q/p ₂,  (16) which yields 1C* _(avg)=1/(2πε₁)ln(g* ₁₂ ²/(g* ₁₁ g* ₂₂)).  (17)

Here, g_(ij) is the average geometrical distance (AGD) between the contours of i-th and j-th conductors,

$\begin{matrix} \begin{matrix} {{\ln\left( g_{ij} \right)} = {\ln\left( {g\left( {P_{i},P_{j}} \right)} \right)}} \\ {{= {\frac{1}{p_{i}p_{j}}{\int_{Pi}\ {{\mathbb{d}{P^{\prime}(X)}}{\int_{P_{j}}{{\ln\left( {{R\left( {X,x} \right)}/R_{0}} \right)}\ {\mathbb{d}P}(x)}}}}}},} \\ {g_{ij}^{*} = {g\left( {P_{i},P_{j}^{*}} \right)}} \\ {{= {g\left( {P_{i}^{*},P_{j}} \right)}},i,{j = 1},2,} \end{matrix} & (18) \end{matrix}$ P_(i)* denotes the mirror image of P_(i) in the e₁/e₂ boundary.

The cross-sections of on-chip coplanar transmission lines as shown in FIGS. 1A and 1B are rectangular, so the right side of Equation 17 is the sum of AGD's between parallel and perpendicular straight line segments, for which simple explicit expressions have been developed.

TABLE 1 Coplanar T-line type (FIGS. 1A & 1B) Single Coupled Signal width, um 9.0 0.70 1.0 0.20 Shield width, um 9.0 0.20 0.40 0.30 Signal-to-shield, 6.0 8.0 4.0  8.0  um Metal thickness, 0.29 0.32 0.55 0.32 um Metal-to-substrate 0.46 1.1 2.1  1.1  um Signal-to-signal, — — 1.2  3.0  um Operation mode — — Common Common Single Single C (−1), 569 72.3 87.2 97.0 83.3 68.6 pF/m(solver) C (−0.5) 167 57.1 81.5 94.4 64.7 59.4 pF/m(solver) C (−0.5), 157 57.7 81.5 92.9 65.1 59.0 pF/m(model) Relative error −5.5% 1.1% 0.02% −1.6% 0.6% −0.8% C (−0.5), model vs. solver

Table 1 compares the values of high frequency capacitance computed using the presented methodology with the results obtained by two-dimensional Ansoft static EM solver (EM2D), for several typical coplanar transmission line structures. In order to get the upper boundary of the error, the case of k₁=−0.5, k₂=−1 is studied. The capacitance C₈=C(−0.5) obtained by the solver is compared with its approximation by Equation 15, while using the solver for C₀=C(−1) and Equation 17 for C*_(avg).

The steps involved in implementing the method of capacitance modeling using the computations detailed above are now described. The aim is to provide a method to calculate the capacitance of a certain two-dimensional structure for specific values of permittivity of the surrounding dielectric material e₁ and the substrate e₂. The method uses extrapolation or interpolation formulae which yield the required solution (referred to as the subject solution) using solutions already computed for the same cross-section geometry with the same value of permittivity of the dielectric e₁ and different value(s) of the permittivity of the substrate e₂ (referred to as the basis solution(s)).

The input information needed for the method of modeling capacitance includes: the permittivity of the surrounding dielectric material e₁, the permittivity of the substrate e₂ of the basis solution(s), and the cross-sectional geometry of the structure. Optionally, the charge density function for the basis solution may be used.

Referring to FIGS. 3A to 3C, flow diagrams 300, 310, 320 are shown of the implementation steps of the method. In FIG. 3A, a flow diagram 300 is shown of the overall method of modeling capacitance in which a subject structure is provided 301 for which the capacitance is to be modeled. The subject structure has a pair of long conductors surrounded by a dielectric material and supported by a substrate. A basis structure is also provided 302 which has the same cross-sectional geometry as the subject structure.

It is assumed 303 that the surrounding dielectric material is a first dielectric with a first permittivity and the substrate is a second dielectric with a second permittivity. At very high frequencies a conductive substrate behaves as a dielectric resulting in this assumption. The first permittivity does not depend on frequency and the second permittivity does depend on frequency. The basis structure has the same first permittivity and a different second permittivity as the subject structure.

At the next step in the flow diagram, the known capacitance is computed 304 for the basis structure. This may be done by using, for example, one of an Electro-Magnetic solver, an analytical formula, a quasi-analytical formula, or lookup tables.

The capacitance for the subject structure can then be modeled 305 for the values of the first and second permittivity based on the known capacitance computed for the basis structure with the same first permittivity and a different second permittivity. The modeling can be carried out using an extrapolation or interpolation formula.

In FIG. 3B, a flow diagram 310 is shown of a method of modeling capacitance with the same initial steps 301, 302, 303 as shown in FIG. 3A. The steps of computing the known capacitance for the basis structure and modeling the capacitance for the subject structure are provided in more detail. At step 314 of FIG. 3B, the reflection coefficients for the subject structure and the basis structure are determined. The mirror capacitance for the basis structure is then determined 315 and the capacitance of the subject structure is computed 316 based on the reflection coefficients and the mirror capacitance.

The steps 314, 315, 316 are shown in more detail in the flow diagram 320 of FIG. 3C.

Referring to FIG. 3C, firstly, the following denotations are defined:

-   -   C₁=the capacitance for the subject solution;     -   C₂=the capacitance for the basis solution;     -   e₂₁=the substrate permittivity e₂ of the subject solution; and     -   e₂₂=the substrate permittivity e₂ of the basis solution.

At the first step 321 of the flow diagram 320, for i=1,2 compute the reflection coefficient, k_(i)=(e₁−e_(2i))/(e₁+e_(2i)). Next, it is determined 322 if the charge density function s(X,k₂) for the basis solution is available. If it is available, compute 323 the “mirror capacitance” C*(k₂)=C*(X₁,X₂,k₂) which is fully defined by s(X,k₂). If the charge density function for the basis solution is not available, compute 324 the “averaged mirror capacitance” C*(k₂)=C*_(avg)(k₂) based on the uniform charge density approximation. Computation of both C*(X₁,x₂, k₂) and C*_(avg)(k₂) is explained above.

In the final step 325, compute C₁ using the following extrapolation formula: C ₁=1/(1/C ₂+(k ₁ −k ₂)/C*(k ₂))

A similar implementation based on interpolation formula can be used which needs no less than two basis solutions.

The proposed approach for high frequency capacitance calculation of on-chip coplanar transmission lines over a substrate can be incorporated into CAD packages for interconnect-aware design and modeling. The approach is free from convergence difficulties and yields expressions which are easy to implement as a computer program. The final explicit semi-analytical expressions are easy to compute, do not contain any fitting parameters and yield reasonable accuracy.

Referring to FIG. 4, a schematic representation of an integrated circuit design system 400 is shown. The system includes a design means 402 which may be implemented by specialized CAD software running on a computer indicated in the figure by a processor 401. The computer provides an interface with a designer.

The design means 402 includes means for modeling capacitance 403 for carrying out the steps of the described method which may be in the form of a computer program product. The means for modeling capacitance 403 includes means for determining 404 the reflection coefficients of the subject and basis structure, means for determining 405 the mirror capacitance of the basis structure and means for computing 406 the capacitance of the subject structure.

The present invention is typically implemented as a computer program product, comprising a set of program instructions for controlling a computer or similar device. These instructions can be supplied preloaded into a system or recorded on a storage medium such as a CD-ROM, or made available for downloading over a network such as the Internet or a mobile telephone network.

Improvements and modifications can be made to the foregoing without departing from the scope of the present invention. Thus variations described for the present invention can be realized in any combination desirable for each particular application. Thus particular limitations, and/or embodiment enhancements described herein, which may have particular advantages to the particular application need not be used for all applications. Also, not all limitations need be implemented in methods, systems and/or apparatus including one or more concepts of the present invention.

The present invention can be realized in hardware, software, or a combination of hardware and software. A visualization tool according to the present invention can be realized in a centralized fashion in one computer system, or in a distributed fashion where different elements are spread across several interconnected computer systems. Any kind of computer system—or other apparatus adapted for carrying out the methods and/or functions described herein—is suitable. A typical combination of hardware and software could be a general purpose computer system with a computer program that, when being loaded and executed, controls the computer system such that it carries out the methods described herein. The present invention can also be embedded in a computer program product, which comprises all the features enabling the implementation of the methods described herein, and which—when loaded in a computer system—is able to carry out these methods.

Computer program means or computer program in the present context include any expression, in any language, code or notation, of a set of instructions intended to cause a system having an information processing capability to perform a particular function either directly or after conversion to another language, code or notation, and/or reproduction in a different material form.

Thus the invention includes an article of manufacture which comprises a computer usable medium having computer readable program code means embodied therein for causing a function described above. The computer readable program code means in the article of manufacture comprises computer readable program code means for causing a computer to effect the steps of a method of this invention. Similarly, the present invention may be implemented as a computer program product comprising a computer usable medium having computer readable program code means embodied therein for causing a function described above. The computer readable program code means in the computer program product comprising computer readable program code means for causing a computer to effect one or more functions of this invention. Furthermore, the present invention may be implemented as a program storage device readable by machine, tangibly embodying a program of instructions executable by the machine to perform method steps for causing one or more functions of this invention.

It is noted that the foregoing has outlined some of the more pertinent objects and embodiments of the present invention. This invention may be used for many applications. Thus, although the description is made for particular arrangements and methods, the intent and concept of the invention is suitable and applicable to other arrangements and applications. It will be clear to those skilled in the art that modifications to the disclosed embodiments can be effected without departing from the spirit and scope of the invention. The described embodiments ought to be construed to be merely illustrative of some of the more prominent features and applications of the invention. Other beneficial results can be realized by applying the disclosed invention in a different manner or modifying the invention in ways known to those familiar with the art. 

1. A method, implemented by a computer device, of modeling capacitance for a subject structure comprising a pair of long conductors surrounded by a dielectric material and supported by a substrate, wherein the capacitance of the conductors is a function of current frequency, the method comprising: receiving, at said computer device, parameters characterizing said surrounding dielectric material of said subject structure as a first dielectric with a first permittivity (ε₁) and the substrate as a second dielectric with a second permittivity (ε₂); providing, at said computer device, a known capacitance (C2) value of a basis structure characterized as having a surrounding dielectric material of same first permittivity (ε₁) and a substrate of a second permittivity (ε₂) different than said second permittivity of said subject structure and having a same cross-sectional geometry of said subject structure; modeling, at said computer device, the capacitance (C₁) of said subject structure for values of the first and second permittivity (ε₁, ε₂) based on known capacitance (C₂) computed for said basis structure with the same first permittivity (ε₁) and a different second permittivity (ε₂), and using said modeled capacitance (C1) for design of an integrated circuit having said subject structure.
 2. A method as claimed in claim 1, wherein the known capacitance is computed for a basis structure with the same cross-section geometry as the subject structure.
 3. A method as claimed in claim 1, wherein the first permittivity does not depend on frequency and the second permittivity does depend on frequency.
 4. A method as claimed in claim 1, wherein the modelled capacitance is high frequency capacitance.
 5. A method as claimed in claim 1, wherein the capacitance of the subject structure is modelled from the capacitance of one or more basis structures using an extrapolation or interpolation formula.
 6. A method as claimed in claim 1, wherein the input information for the method includes: the first permittivity ε₁ of The first dielectric of both the subject structure and the basis structure, the second permittivity ε₂ of The second dielectric of the basis structure, and the cross-sectional geometry of the subject or basis structure.
 7. A method as claimed in claim 1, wherein The known capacitance is provided by one of an Electro-Magnetic solver, an analytical formula, a quasi-analytical formula, and look-up tables.
 8. A method as claimed in claim 1, wherein the surrounding dielectric material is an insulating material and the substrate behaves as a conductor at low frequencies and as a dielectric at very high frequencies.
 9. A method as claimed in claim 1, wherein the dielectric is an oxide dielectric and the substrate is a silicon substrate.
 10. A method as claimed in claim 1, wherein the pair of long conductors each have a width and metal layer thickness of the same order.
 11. A method as claimed in claim 1, wherein the pair of long conductors are spaced apart by a distance of the same order as the width and thickness of the conductors.
 12. A method as claimed in claim 1, wherein the cross-section of each of the long conductors is constant along the length.
 13. A method as claimed in claim 1, wherein each of the long conductors is formed of a single conductor or more than one parallel conductor.
 14. A method as claimed in claim 1, wherein the structure comprises on-chip coplanar transmission lines over a conductive substrate. 